NCP4305 Datasheet by onsemi

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© Semiconductor Components Industries, LLC, 2016
June, 2016 − Rev. 3 1Publication Order Number:
NCP4305/D
NCP4305
Secondary Side
Synchronous Rectification
Driver for High Efficiency
SMPS Topologies
The NCP4305 is high performance driver tailored to control a
synchronous rectification MOSFET in switch mode power supplies.
Thanks to its high performance drivers and versatility, it can be used in
various topologies such as DCM or CCM flyback, quasi resonant
flyback, forward and half bridge resonant LLC.
The combination of externally adjustable minimum off-time and
on-time blanking periods helps to fight the ringing induced by the PCB
layout and other parasitic elements. A reliable and noise less operation
of the SR system is insured due to the Self Synchronization feature. The
NCP4305 also utilizes Kelvin connection of the driver to the MOSFET
to achieve high efficiency operation at full load and utilizes a light load
detection architecture to achieve high efficiency at light load.
The precise turn−off threshold, extremely low turn−off delay time
and high sink current capability of the driver allow the maximum
synchronous rectification MOSFET conduction time and enables
maximum SMPS efficiency. The high accuracy driver and 5 V gate
clamp enables the use of GaN FETs.
Features
Self−Contained Control of Synchronous Rectifier in CCM, DCM and
QR for Flyback, Forward or LLC Applications
Precise True Secondary Zero Current Detection
Typically 12 ns Turn off Delay from Current Sense Input to Driver
Rugged Current Sense Pin (up to 200 V)
Ultrafast Turn−off Trigger Interface/Disable Input (7.5 ns)
Adjustable Minimum ON−Time
Adjustable Minimum OFF-Time with Ringing Detection
Adjustable Maximum ON−Time for CCM Controlling of Primary
QR Controller
Improved Robust Self Synchronization Capability
8 A / 4 A Peak Current Sink / Source Drive Capability
Operating Voltage Range up to VCC = 35 V
Automatic Light−load & Disable Mode
Adaptive Gate Drive Clamp
GaN Transistor Driving Capability (options A and C)
Low Startup and Disable Current Consumption
Maximum Operation Frequency up to 1 MHz
SOIC-8 and DFN−8 (4x4) and WDFN8 (2x2) Packages
These are Pb−Free Devices
Typical Applications
Notebook Adapters
High Power Density AC/DC Power Supplies (Cell
Phone Chargers)
LCD TVs
All SMPS with High Efficiency Requirements
SOIC−8
D SUFFIX
CASE 751
MARKING
DIAGRAMS
4305x = Specific Device Code
x = A, B, C, D or Q
A = Assembly Location
L = Wafer Lot
Y = Year
W = Work Week
M = Date Code
G= Pb−Free Package
1
8
NCP4305x
ALYW G
G
1
8
(Note: Microdot may be in either location)
4305x
ALYWG
G
1
DFN8
MN SUFFIX
CASE 488AF
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See detailed ordering and shipping information on page 49 o
f
this data sheet.
ORDERING INFORMATION
5xMG
G
1
WDFN8
MT SUFFIX
CASE 511AT
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Figure 1. Typical Application Example − LLC Converter with Optional LLD and Trigger Utilization
Figure 2. Typical Application Example − DCM, CCM or QR Flyback Converter with optional LLD and Disabled
TRIG
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Figure 3. Typical Application Example − Primary Side Flyback Converter with optional LLD and Disabled TRIG
Figure 4. Typical Application Example − QR Converter − Capability to Force Primary into CCM Under Heavy
Loads utilizing MAX−TON
ELAPSED EN
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PIN FUNCTION DESCRIPTION
ver. A, B, C, D ver. Q Pin Name Description
1 1 VCC Supply voltage pin
2 2 MIN_TOFF Adjust the minimum off time period by connecting resistor to ground.
3 3 MIN_TON Adjust the minimum on time period by connecting resistor to ground.
4 4 LLD This input modulates the driver clamp level and/or turns the driver off during light load
conditions.
5 TRIG/DIS Ultrafast turn−off input that can be used to turn off the SR MOSFET in CCM applica-
tions in order to improve efficiency. Activates disable mode if pulled−up for more than
100 ms.
6 6 CS Current sense pin detects if the current flows through the SR MOSFET and/or its body
diode. Basic turn−off detection threshold is 0 mV. A resistor in series with this pin can
decrease the turn off threshold if needed.
7 7 GND Ground connection for the SR MOSFET driver and VCC decoupling capacitor. Ground
connection for minimum on and off time adjust resistors, LLD and trigger inputs.
GND pin should be wired directly to the SR MOSFET source terminal/soldering point
using Kelvin connection. DFN8 exposed flag should be connected to GND
8 8 DRV Driver output for the SR MOSFET
5 MAX_TON Adjust the maximum on time period by connecting resistor to ground.
Minimum ON time
generator
MIN_TON
CS
detection
100mA
CS
MIN_TOFF
TRIG/ DISABLE
CS_ON
CS_OFF
DRV
VCC
GND
VCC managment
UVLO
DRV Out
DRIVER
VDD
VDD
CS_RESET
LLD
Disable detection
&
V DRV clamp
modulation
V_DRV
control
ADJ ELAPSED
EN
Minimum OFF
time generator
ADJ
RESET
ELAPSED
10 A Vtrig
Control logic
EN
DISABLE
Disable detection
DISABLE
DISABLE
TRIG
Figure 5. Internal Circuit Architecture − NCP4305A, B, C, D
ELAPSED EN
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Minimum ON time
generator
MIN_TON
CS
detection
100mA
CS
MIN_TOFF
MAX_TON
CS_ON
CS_OFF
DRV
VCC
GND
VCC managment
UVLO
DRV Out
DRIVER
VDD
VDD
CS_RESET
LLD
Disable detection
&
V DRV clamp
modulation
V_DRV
control
ADJ
ELAPSED
EN
Minimum OFF
time generator
ADJ
RESET
ELAPSED
Control logic
EN
DISABLE
DISABLE
ELAPSED
Maximum ON time
generator EN
ADJ
Figure 6. Internal Circuit Architecture − NCP4305Q (CCM QR) with MAX_TON
ABSOLUTE MAXIMUM RATINGS
Rating Symbol Value Unit
Supply Voltage VCC −0.3 to 37.0 V
TRIG/DIS, MIN_TON, MIN_TOFF, MAX_TON, LLD Input Voltage VTRIG/DIS,
VMIN_TON,
VMIN_TOFF,
VMAX_TON, VLLD
−0.3 to VCC V
Driver Output Voltage VDRV −0.3 to 17.0 V
Current Sense Input Voltage VCS −4 to 200 V
Current Sense Dynamic Input Voltage (tPW = 200 ns) VCS_DYN −10 to 200 V
MIN_TON, MIN_TOFF, MAX_TON, LLD, TRIG Input Current IMIN_TON, IMIN_TOFF,
IMAX_TON, ILLD, ITRIG
−10 to 10 mA
Junction to Air Thermal Resistance, 1 oz 1 in2 Copper Area, SOIC8 RqJ−A_SOIC8 160 °C/W
Junction to Air Thermal Resistance, 1 oz 1 in2 Copper Area, DFN8 RqJ−A_DFN8 80 °C/W
Junction to Air Thermal Resistance, 1 oz 1 in2 Copper Area, WDFN8 RqJ−A_WDFN8 160 °C/W
Maximum Junction Temperature TJMAX 150 °C
Storage Temperature TSTG −60 to 150 °C
ESD Capability, Human Body Model, Except Pin 6, per JESD22−A114E ESDHBM 2000 V
ESD Capability, Human Body Model, Pin 6, per JESD22−A114E ESDHBM 1000 V
ESD Capability, Machine Model, per JESD22−A115−A ESDMM 200 V
ESD Capability, Charged Device Model, Except Pin 6, per JESD22−C101F ESDCDM 750 V
ESD Capability, Charged Device Model, Pin 6, per JESD22−C101F ESDCDM 250 V
Stresses exceeding those listed in the Maximum Ratings table may damage the device. If any of these limits are exceeded, device functionality
should not be assumed, damage may occur and reliability may be affected.
1. This device meets latch−up tests defined by JEDEC Standard JESD78D Class I.
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RECOMMENDED OPERATING CONDITIONS
Parameter Symbol Min Max Unit
Maximum Operating Input Voltage VCC 35 V
Operating Junction Temperature TJ−40 125 °C
Functional operation above the stresses listed in the Recommended Operating Ranges is not implied. Extended exposure to stresses beyond
the Recommended Operating Ranges limits may affect device reliability.
ELECTRICAL CHARACTERISTICS
−40°C TJ 125°C; VCC = 12 V; CDRV = 0 nF; RMIN_TON = RMIN_TOFF = 10 kW; VTRIG/DIS = 0 V; VLLD = 0 V; VCS = −1 to +4 V; fCS =
100 kHz, DCCS = 50%, unless otherwise noted. Typical values are at TJ = +25°C
Parameter Test Conditions Symbol Min Typ Max Unit
SUPPLY SECTION
VCC UVLO (ver. B & C) VCC rising VCCON 8.3 8.8 9.4 V
VCC falling VCCOFF 7.3 7.8 8.3
VCC UVLO Hysteresis (ver. B & C) VCCHYS 1.0 V
VCC UVLO (ver. A, D & Q) VCC rising VCCON 4.20 4.45 4.80 V
VCC falling VCCOFF 3.70 3.95 4.20
VCC UVLO Hysteresis
(ver. A, D & Q) VCCHYS 0.5 V
Start−up Delay VCC rising from 0 to VCCON + 1 V @ tr = 10 mstSTART_DEL 75 125 ms
Current Consumption,
RMIN_TON = RMIN_TOFF = 0 kW
CLOAD = 0 nF, fSW = 500 kHz A, C ICC 3.3 4.0 5.6 mA
B, D, Q 3.8 4.5 6.0
CLOAD = 0 nF, fSW = 500 kHz,
WDFN A, C 3.0 4.0 5.6
B, D, Q 3.5 4.5 6.0
CLOAD = 1 nF, fSW = 500 kHz A, C 4.5 6.0 7.5
B, D, Q 7.7 9.0 10.7
CLOAD = 10 nF, fSW = 500 kHz A, C 20 25 30
B, D, Q 40 50 60
Current Consumption No switching, VCS = 0 V,
RMIN_TON = RMIN_TOFF = 0 k ICC 1.5 2.0 2.5 mA
Current Consumption below UVLO No switching, VCC = VCCOFF – 0.1 V, VCS = 0 V ICC_UVLO 75 125 mA
Current Consumption in Disable
Mode VLLD = VCC − 0.1 V, VCS = 0 V ICC_DIS 40 55 70 mA
VTRIG = 5 V, VLLD = VCC – 3 V, VCS = 0 V 45 65 80
DRIVER OUTPUT
Output Voltage Rise−Time CLOAD = 10 nF, 10% to 90% VDRVMAX tr40 55 ns
Output Voltage Fall−Time CLOAD = 10 nF, 90% to 10% VDRVMAX tf20 35 ns
Driver Source Resistance RDRV_SOURCE 1.2 W
Driver Sink Resistance RDRV_SINK 0.5 W
Output Peak Source Current IDRV_SOURCE 4 A
Output Peak Sink Current IDRV_SINK 8 A
Maximum Driver Output Voltage VCC = 35 V, CLOAD > 1 nF, VLLD = 0 V,
(ver. B, D and Q) VDRVMAX 9.0 9.5 10.5 V
VCC = 35 V, CLOAD > 1 nF, VLLD = 0 V, (ver. A, C) 4.3 4.7 5.5
Minimum Driver Output Voltage VCC = VCCOFF + 200 mV, VLLD = 0 V, (ver. B) VDRVMIN 7.2 7.8 8.5 V
VCC = VCCOFF + 200 mV, VLLD = 0 V, (ver. C) 4.2 4.7 5.3
VCC = VCCOFF + 200 mV, VLLD = 0 V,
(ver. A, D, Q) 3.6 4.0 4.4
Minimum Driver Output Voltage VLLD = VCC − VLLDREC V VDRVLLDMIN 0.0 0.4 1.2 V
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ELECTRICAL CHARACTERISTICS
−40°C TJ 125°C; VCC = 12 V; CDRV = 0 nF; RMIN_TON = RMIN_TOFF = 10 kW; VTRIG/DIS = 0 V; VLLD = 0 V; VCS = −1 to +4 V; fCS =
100 kHz, DCCS = 50%, unless otherwise noted. Typical values are at TJ = +25°C
Parameter UnitMaxTypMinSymbolTest Conditions
CS INPUT
Total Propagation Delay From CS
to DRV Output On VCS goes down from 4 to −1 V, tf_CS = 5 ns tPD_ON 35 60 ns
Total Propagation Delay From CS
to DRV Output Off VCS goes up from −1 to 4 V, tr_CS = 5 ns tPD_OFF 12 23 ns
CS Bias Current VCS = −20 mV ICS −105 −100 −95 mA
Turn On CS Threshold Voltage VTH_CS_ON −120 −75 −40 mV
Turn Off CS Threshold Voltage Guaranteed by Design VTH_CS_OFF −1 0 mV
Turn Off Timer Reset Threshold
Voltage VTH_CS_RESET 0.42 0.48 0.54 V
CS Leakage Current VCS = 200 V ICS_LEAKAGE 0.4 mA
TRIGGER DISABLE INPUT
Minimum Trigger Pulse Duration VTRIG = 5 V; Shorter pulses may not be
proceeded tTRIG_PW_MIN 10 ns
Trigger Threshold Voltage VTRIG_TH 1.87 2.02 2.18 V
Trigger to DRV Propagation Delay VTRIG goes from 0 to 5 V, tr_TRIG = 5 ns tPD_TRIG 7.5 12.5 ns
Trigger Blank Time After DRV
Turn−on Event VCS drops below VTH_CS_ON tTRIG_BLANK 35 50 65 ns
Delay to Disable Mode VTRIG = 5 V tDIS_TIM 75 100 125 ms
Disable Recovery Timer VTRIG goes down from 5 to 0 V tDIS_REC 5 8 13 ms
Minimum Pulse Duration to Disable
Mode End VTRIG = 0 V; Shorter pulses may not be
proceeded tDIS_END_MIN 200 ns
Pull Down Current VTRIG = 5 V ITRIG 9 13 16 mA
MINIMUM tON and tOFF ADJUST
Minimum tON time RMIN_TON = 0 WtON_MIN 35 55 75 ns
Minimum tOFF time RMIN_TOFF = 0 WtOFF_MIN 190 245 290 ns
Minimum tON time RMIN_TON = 10 kWtON_MIN 0.92 1.00 1.08 ms
Minimum tOFF time RMIN_TOFF = 10 kWtOFF_MIN 0.92 1.00 1.08 ms
Minimum tON time RMIN_TON = 50 kWtON_MIN 4.62 5.00 5.38 ms
Minimum tOFF time RMIN_TOFF = 50 kWtOFF_MIN 4.62 5.00 5.38 ms
MAXIMUM tON ADJUST
Maximum tON Time VMAX_TON = 3 V tON_MAX 4.3 4.8 5.3 ms
Maximum tON Time VMAX_TON = 0.3 V tON_MAX 41 48 55 ms
Maximum tON Output Current VMAX_TON = 0.3 V IMAX_TON −105 −100 −95 mA
LLD INPUT
Disable Threshold VLLD_DIS = VCC − VLLD VLLD_DIS 0.8 0.9 1.0 V
Recovery Threshold VLLD_REC = VCC − VLLD VLLD_REC 0.9 1.0 1.1 V
Disable Hysteresis VLLD_DISH 0.1 V
Disable Time Hysteresis Disable to Normal, Normal to Disable tLLD_DISH 45 ms
Disable Recovery Time tLLD_DIS_REC 7.0 12.5 16.0 ms
Low Pass Filter Frequency fLPLLD 6 10 13 kHz
Driver Voltage Clamp Threshold VDRV = VDRVMAX, VLLDMAX = VCC − VLLD VLLDMAX 2.0 V
Product parametric performance is indicated in the Electrical Characteristics for the listed test conditions, unless otherwise noted. Product
performance may not be indicated by the Electrical Characteristics if operated under different conditions.
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TYPICAL CHARACTERISTICS
Figure 7. VCCON and VCCOFF Levels,
ver. A, D, Q Figure 8. VCCON and VCCOFF Levels,
ver. B, C
TJ (°C) TJ (°C)
100806040200−20−40
3.7
3.8
3.9
4.1
4.2
4.4
4.6
4.7
100806040200−20−40
7.3
7.5
7.7
8.1
8.3
8.7
8.9
9.3
VCC (V)
VCC (V)
120
4.0
4.3
4.5 VCCON
VCCOFF
VCCON
VCCOFF
120
7.9
8.5
9.1
DRV DRV DRV
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TYPICAL CHARACTERISTICS
Figure 9. Current Consumption, CDRV = 0 nF,
fCS = 500 kHz, ver. D Figure 10. Current Consumption, VCC =
VCCOFF − 0.1 V, VCS = 0 V, ver. D
VCC (V) TJ (°C)
3025 3520151050
0
1
2
3
4
5
6
1201006040200−20−40
0
20
40
60
80
100
120
Figure 11. Current Consumption, VCC = 12 V,
VCS = −1 to 4 V, fCS = 500 kHz, ver. A Figure 12. Current Consumption, VCC = 12 V,
VCS = −1 to 4 V, fCS = 500 kHz, ver. D
TJ (°C) TJ (°C)
100806040200−20−40
0
5
10
15
20
25
30
100806040200−20−40
0
10
20
30
40
50
60
Figure 13. Current Consumption in Disable,
VCC = 12 V, VCS = 0 V, VLLD = VCC − 0.1 V Figure 14. Current Consumption in Disable,
VCC = 12 V, VCS = 0 V, VLLD = VCC − 3 V, VTRIG =
5 V
TJ (°C) TJ (°C)
100806040200−20−40
40
45
50
55
60
65
70
100806040200−20−40
45
50
55
60
65
70
75
80
ICC (mA)
ICC_UVLO (mA)
ICC (mA)
ICC (mA)
ICC_DIS (mA)
ICC_DIS (mA)
TJ = 85°CTJ = 55°CTJ = 125°C
TJ = 25°C
TJ = 0°C
TJ = −20°C
TJ = −40°C
80
120
CDRV = 0 nF
CDRV = 1 nF
CDRV = 10 nF
CDRV = 0 nF
CDRV = 1 nF
CDRV = 10 nF
120
120 120
// //
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TYPICAL CHARACTERISTICS
Figure 15. CS Current, VCS = −20 mV Figure 16. CS Current, VCC = 12 V
TJ (°C) VCS (V)
100806040200−20−40
−110
−106
−104
−100
−98
−96
−94
−90
0.80.60.20−0.2−0.4−0.8−1.0
−1.4
−1.2
−1.0
−0.8
−0.6
−0.4
−0.2
0
Figure 17. Supply Current vs. CS Voltage,
VCC = 12 V Figure 18. CS Turn−on Threshold
VCS (V) TJ (°C)
3210−1−2−3−4
0
0.5
1.0
1.5
2.0
2.5
3.0
100806040200−20−40
−150
−130
−110
−90
−70
−50
−30
Figure 19. CS Turn−off Threshold Figure 20. CS Reset Threshold
TJ (°C) TJ (°C)
100806040200−20−40
−2.0
−1.5
−1.0
−0.5
0
0.5
1.0
0.40
0.45
0.50
0.55
0.60
ICS (mA)
ICS (mA)
ICC (mA)
VTH_CS_ON (mV)
VTH_CS_OFF (mV)
VTH_CS_RESET (V)
120
−92
−102
−108
−0.6 0.4 1.0
4
TJ = 125°C
TJ = 85°C
TJ = 55°C
TJ = 25°C
TJ = 0°C
TJ = −20°C
TJ = −40°C
TJ = 125°C
TJ = 85°C
TJ = 55°C
TJ = 25°C
TJ = 0°C
TJ = −20°C
TJ = −40°C
120
120 100806040200−20−40 120
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TYPICAL CHARACTERISTICS
Figure 21. CS Reset Threshold Figure 22. CS Leakage, VCS = 200 V
VCC (V) TJ (°C)
302520 35151050
0.30
0.35
0.45
0.50
0.60
0.65
0.70
0.80
100 1206040200−20−40
0
20
60
80
120
140
180
200
Figure 23. Propagation Delay from CS to DRV
Output On Figure 24. Propagation Delay from CS to DRV
Output Off
TJ (°C) TJ (°C)
100806040200−20−40
20
25
30
35
40
50
55
60
100806040200−20−40
4
6
10
12
16
18
22
24
Figure 25. Trigger Threshold, VCC = 12 V Figure 26. Trigger Threshold
TJ (°C) VCC (V)
100806040200−20−40
1.95
1.97
2.01
2.03
2.07
2.09
2.11
2.15
3025 3520151050
1.5
1.6
1.8
1.9
2.0
2.2
2.4
2.5
VTH_CS_RESET (V)
ICS_LEAKAGE (nA)
tPD_ON (ns)
tPD_OFF (ns)
VTRIG_TH (V)
VTRIG_TH (V)
0.40
0.55
0.75
80
40
100
160
120
45
120
8
14
20
120
1.99
2.05
2.13
1.7
2.1
2.3
TJ = 125°C
TJ = 85°C
TJ = 55°C
TJ = 25°C
TJ = 0°C
TJ = −20°C
TJ = −40°C
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TYPICAL CHARACTERISTICS
Figure 27. Trigger Pull Down Current Figure 28. Trigger Pull Down Current,
VCC = 12 V
TJ (°C) VTRIG (V)
100806040200−20−40
9
10
11
12
13
14
15
16
4.54.03.02.52.01.00.50
0
2
4
6
8
10
12
14
Figure 29. Propagation Delay from Trigger to
Driver Output Off Figure 30. Delay to Disable Mode, VTRIG = 5 V
TJ (°C) TJ (°C)
100806040200−20−40
2
4
6
8
10
12
14
100806040200−20−40
85
90
95
100
105
110
115
Figure 31. Minimum On−time RMIN_TON = 0 WFigure 32. Minimum On−time RMIN_TON = 10 kW
TJ (°C) TJ (°C)
100806040200−20−40
35
40
45
50
55
60
70
75
100806040200−20−40
0.92
0.94
0.96
0.98
1.00
1.04
1.06
1.08
ITRIG (mA)
ITRIG (mA)
tPD_TRIG (ns)
tDIS_TIM (ms)
tMIN_TON (ns)
tMIN_TON (ms)
120 1.5 3.5 5.0
TJ = 125°C
TJ = 85°C
TJ = 55°C
TJ = 25°C
TJ = 0°C
TJ = −20°C
TJ = −40°C
120 120
120
65
120
1.02
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TYPICAL CHARACTERISTICS
Figure 33. Minimum On−time RMIN_TON = 50 kWFigure 34. Minimum Off−time RMIN_TOFF = 0 W
TJ (°C) TJ (°C)
100806040200−20−40
4.6
4.7
4.8
4.9
5.0
5.2
5.3
5.4
100806040200−20−40
190
200
220
230
240
260
270
290
Figure 35. Minimum Off−time RMIN_TOFF =
10 kW
Figure 36. Minimum Off−time RMIN_TOFF =
50 kW
TJ (°C) TJ (°C)
100806040200−20−40
0.92
0.94
0.96
1.00
1.02
1.04
1.06
1.08
100806040200−20−40
4.6
4.7
4.8
4.9
5.0
5.1
5.3
5.4
Figure 37. Minimum On−time RMIN_TON = 10 kWFigure 38. Minimum Off−time RMIN_TOFF =
10 kW
VCC (V) VCC (V)
302520 35151050
0.92
0.94
0.96
0.98
1.00
1.02
1.03
1.04
35302520151050
092
0.94
0.96
0.98
1.00
1.02
1.06
1.08
tMIN_TON (ms)
tMIN_TOFF (ns)
tMIN_TOFF (ms)
tMIN_TOFF (ms)
tMIN_TON (ms)
tMIN_TOFF (ms)
120
5.1
120
210
250
280
120
0.98
120
5.2
1.01
1.04
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TYPICAL CHARACTERISTICS
Figure 39. Driver and Output Voltage, ver. B, D
and Q Figure 40. Driver Output Voltage, ver. A and C
TJ (°C) TJ (°C)
100806040200−20−40
9.0
9.2
9.4
9.6
9.8
10.0
10.2
10.4
100806040200−20−40
4.3
4.5
4.7
4.9
5.1
5.3
5.5
Figure 41. Maximum On−time, ver. Q Figure 42. Maximum On−time, VMAX_TON = 3 V,
ver. Q
VMAX_TON (V) TJ (°C)
3.02.52.01.51.00.50
0
5
15
20
25
35
45
50
100806040200−20−40
4.3
4.4
4.6
4.7
4.8
5.0
5.1
5.3
Figure 43. Maximum On−time, VMAX_TON =
0.3 V, ver. Q
TJ (°C)
100806040200−20−40
41
43
45
47
49
51
53
55
VDRV (V)
VDRV (V)
tMAX_TON (ms)
tMAX_TON (ms)
tMAX_TON (ms)
120
VCC = 12 V, CDRV = 0 nF
VCC = 12 V, CDRV = 1 nF
VCC = 12 V, CDRV = 10 nF
VCC = 35 V, CDRV = 0 nF
VCC = 35 V, CDRV = 1 nF
VCC = 35 V, CDRV = 10 nF
VCC = 12 V, CDRV = 0 nF
VCC = 12 V, CDRV = 1 nF
VCC = 12 V, CDRV = 10 nF
VCC = 35 V, CDRV = 0 nF
VCC = 35 V, CDRV = 1 nF
VCC = 35 V, CDRV = 10 nF
120
TJ = 125°C
TJ = 85°C
TJ = 55°C
TJ = 25°C
TJ = 0°C
TJ = −20°C
TJ = −40°C
10
30
40
120
4.5
4.9
5.2
120
NCP4305
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15
APPLICATION INFORMATION
General description
The NCP4305 is designed to operate either as a standalone
IC or as a companion IC to a primary side controller to help
achieve efficient synchronous rectification in switch mode
power supplies. This controller features a high current gate
driver along with high−speed logic circuitry to provide
appropriately timed drive signals to a synchronous
rectification MOSFET. With its novel architecture, the
NCP4305 has enough versatility to keep the synchronous
rectification system efficient under any operating mode.
The NCP4305 works from an available voltage with range
from 4 V (A, D & Q options) or 8 V (B & C options) to 35 V
(typical). The wide VCC range allows direct connection to
the SMPS output voltage of most adapters such as
notebooks, cell phone chargers and LCD TV adapters.
Precise turn-off threshold of the current sense comparator
together with an accurate offset current source allows the
user to adjust for any required turn-off current threshold of
the SR MOSFET switch using a single resistor. Compared
to other SR controllers that provide turn-off thresholds in the
range of −10 mV to −5 mV, the NCP4305 offers a turn-off
threshold of 0 mV. When using a low RDS(on) SR (1 mW)
MOSFET our competition, with a −10 mV turn off, will turn
off with 10 A still flowing through the SR FET, while our
0 mV turn off turns off the FET at 0 A; significantly
reducing the turn-off current threshold and improving
efficiency. Many of the competitor parts maintain a drain
source voltage across the MOSFET causing the SR
MOSFET to operate in the linear region to reduce turn−off
time. Thanks to the 8 A sink current of the NCP4305
significantly reduces turn off time allowing for a minimal
drain source voltage to be utilized and efficiency
maximized.
To overcome false triggering issues after turn-on and
turn−off events, the NCP4305 provides adjustable minimum
on-time and off-time blanking periods. Blanking times can
be adjusted independently of IC VCC using external
resistors connected to GND. If needed, blanking periods can
be modulated using additional components.
An extremely fast turn−off comparator, implemented on
the current sense pin, allows for NCP4305 implementation
in CCM applications without any additional components or
external triggering.
An ultrafast trigger input offers the possibility to further
increase efficiency of synchronous rectification systems
operated in CCM mode (for example, CCM flyback or
forward). The time delay from trigger input to driver turn off
event is tPD_TRIG. Additionally, the trigger input can be used
to disable the IC and activate a low consumption standby
mode. This feature can be used to decrease standby
consumption of an SMPS. If the trigger input is not wanted
than the trigger pin can be tied to GND or an option can be
chosen to replace this pin with a MAX_TON input.
An output driver features capability to keep SR transistor
closed even when there is no supply voltage for NCP4305.
SR transistor drain voltage goes up and down during SMPS
operation and this is transferred through drain gate
capacitance to gate and may turn on transistor. NCP4305
uses this pulsing voltage at SR transistor gate (DRV pin) and
uses it internally to provide enough supply to activate
internal driver sink transistor. DRV voltage is pulled low
(not to zero) thanks to this feature and eliminate the risk of
turned on SR transistor before enough VCC is applied to
NCP4305.
Some IC versions include a MAX_TON circuit that helps
a quasi resonant (QR) controller to work in CCM mode
when a heavy load is present like in the example of a
printers motor starting up.
Finally, the NCP4305 features a special pin (LLD) that
can be used to reduce gate driver voltage clamp according
to application load conditions. This feature helps to reduce
issues with transition from disabled driver to full driver
output voltage and back. Disable state can be also activated
through this pin to decrease power consumption in no load
conditions. If the LLD feature is not wanted then the LLD
pin can be tied to GND.
Current Sense Input
Figure 44 shows the internal connection of the CS
circuitry on the current sense input. When the voltage on the
secondary winding of the SMPS reverses, the body diode of
M1 starts to conduct current and the voltage of M1’s drain
drops approximately to −1 V. The CS pin sources current of
100 mA that creates a voltage drop on the RSHIFT_CS resistor
(resistor is optional, we recommend shorting this resistor).
Once the voltage on the CS pin is lower than VTH_CS_ON
threshold, M1 is turned−on. Because of parasitic
impedances, significant ringing can occur in the application.
To overcome false sudden turn−off due to mentioned
ringing, the minimum conduction time of the SR MOSFET
is activated. Minimum conduction time can be adjusted
using the RMIN_TON resistor.
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Figure 44. Current Sensing Circuitry Functionality
The SR MOSFET is turned-off as soon as the voltage on
the CS pin is higher than VTH_CS_OFF (typically −0.5 mV
minus any voltage dropped on the optional RSHIFT_CS). For
the same ringing reason, a minimum off-time timer is
asserted once the VCS goes above VTH_CS_RESET. The
minimum off-time can be externally adjusted using
RMIN_TOFF resistor. The minimum off−time generator can
be re−triggered by MIN_TOFF reset comparator if some
spurious ringing occurs on the CS input after SR MOSFET
turn−off event. This feature significantly simplifies SR
system implementation in flyback converters.
In an LLC converter the SR MOSFET M1 channel
conducts while secondary side current is decreasing (refer to
Figure 45). Therefore the turn−off current depends on
MOSFET RDSON. The −0.5 mV threshold provides an
optimum switching period usage while keeping enough time
margin for the gate turn-off. The RSHIFT_CS resistor
provides the designer with the possibility to modify
(increase) the actual turn−on and turn−off secondary current
thresholds. To ensure proper switching, the min_tOFF timer
is reset, when the VDS of the MOSFET rings and falls down
past the VTH_CS_RESET. The minimum off−time needs to
expire before another drive pulse can be initiated. Minimum
off−time timer is started again when VDS rises above
VTH_CS_RESET.
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VDS =V
CS
VTH_CS_RESET –(R
SHIFT_CS*ICS )
VTH_CS_OFF–(R
SHIFT_CS*ICS )
VTH_CS_ON–(R
SHIFT_CS*ICS )
VDRV
Min ONtime
t
Min OFFtime
Min tOFF timer was
stopped here because
of VCS<VTH_CS_RESET
tMIN_TON
tMIN_TOFF
ISEC
The tMIN_TON and tMIN_TOFF are adjustable by RMIN_TON and RMIN_TOFF resistors
Turn−on delay Turnoff delay
Figure 45. CS Input Comparators Thresholds and Blanking Periods Timing in LLC
VDS =V
CS
VTH_CS_RESET –(R
SHIFT_CS*ICS)
VTH_CS_OFF–(R
SHIFT_CS*ICS)
VTH_CS_ON–(R
SHIFT_CS*ICS)
VDRV
Min ONtime
t
Min OFFtime
tMIN_TON
tMIN_TOFF
ISEC
The tMIN_TON and tMIN_TOFF are adjustable by RMIN_TON and RMIN_TOFF resistors
Turn−on delay Turnoff delay
Min tOFF timer was
stopped here because
of VCS<VTH_CS_RESET
Figure 46. CS Input Comparators Thresholds and Blanking Periods Timing in Flyback
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If no RSHIFT_CS resistor is used, the turn-on, turn-off and
VTH_CS_RESET thresholds are fully given by the CS input
specification (please refer to electrical characteristics table).
The CS pin offset current causes a voltage drop that is equal
to:
VRSHIFT_CS +RSHIFT_CS *I
CS (eq. 1)
Final turn−on and turn off thresholds can be then calculated
as:
VCS_TURN_ON +VTH_CS_ON *ǒRSHIFT_CS *I
CSǓ(eq. 2)
VCS_TURN_OFF +VTH_CS_OFF *ǒRSHIFT_CS *I
CSǓ(eq. 3)
VCS_RESET +VTH_CS_RESET *ǒRSHIFT_CS *I
CSǓ(eq. 4)
Note that RSHIFT_CS impact on turn-on and VTH_CS_RESET
thresholds is less critical than its effect on the turn−off
threshold.
It should be noted that when using a SR MOSFET in a
through hole package the parasitic inductance of the
MOSFET package leads (refer to Figure 47) causes a
turn−off current threshold increase. The current that flows
through the SR MOSFET experiences a high Di(t)/Dt that
induces an error voltage on the SR MOSFET leads due to
their parasitic inductance. This error voltage is proportional
to the derivative of the SR MOSFET current; and shifts the
CS input voltage to zero when significant current still flows
through the MOSFET channel. As a result, the SR MOSFET
is turned−off prematurely and the efficiency of the SMPS is
not optimized − refer to Figure 48 for a better understanding.
Figure 47. SR System Connection Including MOSFET and Layout Parasitic Inductances in LLC Application
cm mm are snv cm 2% :2 cm mm 9 chz
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Figure 48. Waveforms From SR System Implemented in LLC Application and Using MOSFET in TO220 Package
With Long Leads − SR MOSFET channel Conduction Time is Reduced
Note that the efficiency impact caused by the error voltage
due to the parasitic inductance increases with lower
MOSFETs RDS(on) and/or higher operating frequency.
It is thus beneficial to minimize SR MOSFET package
leads length in order to maximize application efficiency. The
optimum solution for applications with high secondary
current Di/Dt and high operating frequency is to use
lead−less SR MOSFET i.e. SR MOSFET in SMT package.
The parasitic inductance of a SMT package is negligible
causing insignificant CS turn−off threshold shift and thus
minimum impact to efficiency (refer to Figure 49).
nun currm I Pvmary Currant cm mm m2 sav ch: 2m 9 and IUDA 9 Ghz
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Figure 49. Waveforms from SR System Implemented in LLC Application and Using MOSFET in SMT Package with
Minimized Parasitic Inductance − SR MOSFET Channel Conduction Time is Optimized
It can be deduced from the above paragraphs on the
induced error voltage and parameter tables that turn−off
threshold precision is quite critical. If we consider a SR
MOSFET with RDS(on) of 1 mW, the 1 mV error voltage on
the CS pin results in a 1 A turn-off current threshold
difference; thus the PCB layout is very critical when
implementing the SR system. Note that the CS turn-off
comparator is referred to the GND pin. Any parasitic
impedance (resistive or inductive − even on the magnitude
of mW and nH values) can cause a high error voltage that is
then evaluated by the CS comparator. Ideally the CS
turn−off comparator should detect voltage that is caused by
secondary current directly on the SR MOSFET channel
resistance. In reality there will be small parasitic impedance
on the CS path due to the bonding wires, leads and soldering.
To assure the best efficiency results, a Kelvin connection of
the SR controller to the power circuitry should be
implemented. The GND pin should be connected to the SR
MOSFET source soldering point and current sense pin
should be connected to the SR MOSFET drain soldering
point − refer to Figure 47. Using a Kelvin connection will
avoid any impact of PCB layout parasitic elements on the SR
controller functionality; SR MOSFET parasitic elements
will still play a role in attaining an error voltage. Figure 50
and Figure 51 show examples of SR system layouts using
MOSFETs in TO220 and SMT packages. It is evident that
the MOSFET leads should be as short as possible to
minimize parasitic inductances when using packages with
leads (like TO220). Figure 51 shows how to layout design
with two SR MOSFETs in parallel. It has to be noted that it
is not easy task and designer has to paid lot of attention to do
symmetric Kelvin connection.
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Figure 50. Recommended Layout When Using SR
MOSFET in TO220 Package
Figure 51. Recommended Layout When Using SR
MOSFET in SMT Package (2x SO8 FL)
Trigger/Disable input
The NCP4305 features an ultrafast trigger input that
exhibits a maximum of tPD_TRIG delay from its activation to
the start of SR MOSFET turn−off of process. This input can
be used in applications operated in deep Continues
Conduction Mode (CCM) to further increase efficiency
and/or to activate disable mode of the SR driver in which the
consumption of the NCP4305 is reduced to maximum of
ICC_DIS.
NCP4305 is capable to turn−off the SR MOSFET reliably
in CCM applications just based on CS pin information only,
without using the trigger input. However, natural delay of
the ZCD comparator and DRV turn−off delay increase
overlap between primary and secondary MOSFETs
switching (also known as cross conduction). If one wants to
achieve absolutely maximum efficiency with deep CCM
applications, then the trigger signal coming from the
primary side should be applied to the trigger pin. The trigger
input then turns the SR MOSFET off slightly before the
secondary winding voltage reverses. There are several
possibilities for transferring the trigger signal from the
primary to the secondary side − refer to Figures 66 and 67.
The trigger signal is blanked for tTRIGBLANK after the
DRV turn−on process has begun. The blanking technique is
used to increase trigger input noise immunity against the
parasitic ringing that is present during the turn on process
due to the SMPS layout. The trigger input is supersedes the
CS input except trigger blanking period. TRIG/DIS signal
turns the SR MOSFET off or prohibits its turn−on when the
Trigger/Disable pin is pulled above VTRIG_TH.
The SR controller enters disable mode when the trigger
pin is pulled−up for more than tDIS_TIM. In disable mode the
IC consumption is significantly reduced. To recover from
disable mode and enter normal operation, the TRIG/DIS pin
is pulled low at least for tDIS_END.
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VDS =V
CS
VTH _CS _RESET
VTH _CS _OFF
VTH _CS _ON
VTRIG/DIS
VDRV
t
t1 t2 t3 t4 t5 t6 t7 t8 t9
Figure 52. Trigger Input Functionality Waveforms Using the Trigger to Turn−off and Block the DRV Signal
Figure 52 shows basic Trigger/Disable input
functionality. At t1 the Trigger/Disable pin is pulled low to
enter into normal operation. At t2 the CS pin is dropped
below the VTH_CS_ON, signaling to the NCP4305 to start to
turn the SR MOSFET on. At t3 the NCP4305 begins to drive
the MOSFET. At t4, the SR MOSFET is conducting and the
Trigger/Disable pin is pulled high. This high signal on the
Trigger/Disable pin almost immediately turns off the drive
to the SR MOSFET, turning off the MOSFET. The DRV is
not turned−on in other case (t6) because the trigger pin is
high in the time when CS pin signal crosses turn−on
threshold. This figure clearly shows that the DRV can be
asserted only on falling edge of the CS pin signal in case the
trigger input is at low level (t2).
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VDS =V
CS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
TRIG/DIS blank
Min ONtime
VDRV
tTRIGBLANK
t
t1 t2 t3
Figure 53. Trigger Input Functionality Waveforms − Trigger Blanking
In Figure 53 above, at time t1 the CS pin falls below the
VTH_CS_ON while the Trigger is low setting in motion the
DRV signal that appears at t2. At time t2 the DRV signal and
Trigger blanking clock begin. Trigger/Disable signal goes
high shortly after time t2. Due to the Trigger blanking clock
(tTRIG_BLANK) the Triggers high signal does not affect the
DRV signal until the tTRIGBLANK timer has expired. At time
t3 the Trigger/Disable signal is re evaluated and the DRV
signal is turned off. The TRIG/DIS input is blanked for
tTRIGBLANK after DRV set signal to avoid undesirable
behavior during SR MOSFET turn−on event. The blanking
time in combination with high threshold voltage
(VTRIG_TH) prevent triggering on ringing and spikes that are
present on the TRIG/DIS input pin during the SR MOSFET
turn−on process. Controllers response to the narrow pulse
on the Trigger/Disable pin is depicted in Figure 53 − this
short trigger pulse enables to turn the DRV on for
tTRIG_BLANK. Note that this case is valid only if device not
entered disable mode before.
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VDS =V
CS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
TRIG/DIS blank
MIN ONTIME
VDRV
tTRIGBLANK
t
t0 t1 t2 t3 t4 t5 t6
Figure 54. Trigger Input Functionality Waveforms − Trigger Blanking Acts Like a Filter
Figure 54 above shows almost the same situation as in
Figure 53 with one main exception; the TRIG/DIS signal
was not high after trigger blanking timer expired so the DRV
signal remains high. The advantage of the trigger blanking
time during DRV turn−on is evident from Figure 54 since it
acts like a filter on the Trigger/Disable pin. Rising edge of
the DRV signal may cause spikes on the trigger input. If it
wasn’t for the TRIG/DIS blanking these spikes, in
combination with ultra−fast performance of the trigger
logic, could turn the SR MOSFET off in an inappropriate
time.
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VDS =V
CS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
Min ON−time
VDRV
t
t0 t1 t2 t3 t4 t6 t7 t8t5
Figure 55. Trigger Input Functionality Waveforms − Trigger Over Ride, CS Turn Off and Min On−time
Figure 55 depicts all possible driver turn−off events in
details when correct VCC is applied. Controller driver is
disabled based on trigger input signal in time t2; the trigger
input overrides the minimum on−time period.
Driver is turned−off according to the CS (VDS) signal (t5
marker) and when minimum on−time period elapsed
already. TRIG/DIS signal needs to be LOW during this
event.
If the CS (VDS) voltage reaches VTH_CS_OFF threshold
before minimum on−time period ends (t7) and the
Trigger/Disable pin is low the DRV is turned−off on the
falling edge of the minimum on−time period (t8 time marker
in Figure 55). This demonstrates the fact that the Trigger
over rides the minimum on−time. Minimum on−time has
higher priority than the CS signal.
In Figure 56 the trigger input is low the whole time and the
DRV pulses are purely a function of the CS signal and the
minimum on−time. The first DRV pulse terminated based on
the CS signal and another two DRV pulses are prolonged till
the minimum on−time period end despite the CS signal
crosses the VTH_CS_OFF threshold earlier.
If a minimum on−time is too long the situation that occurs
after time marker t6 Figure 56 can occur, is not correct and
should be avoided. The minimum tON period should be
selected shorter to overcome situation that the SR MOSFET
is turned−on for too long time. The secondary current then
changes direction and energy flows back to the transformer
that result in reduced application efficiency and also in
excessive ringing on the primary and secondary MOSFETs.
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VDS =V
CS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
Min ONtime
VDRV
t
t0 t1 t2 t3 t4 t6 t7 t8t5 t9
Figure 56. Minimum On−Time Priority
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VDS =V
CS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
VDRV
t
t0 t1 t2 t3 t4
Min OFFtime
t5
Min ON−time
t6 t7 t8 t9 t10
Figure 57. Trigger Input Functionality Waveforms − Two Pulses at One Cycle
Figure 57 shows IC behavior in case the trigger signal
features two pulses during one cycle of the VDS (CS) signal.
The trigger goes low enables the DRV just before time t1 and
DRV turns−on because the VDS voltage drops under
VTH_CS_ON threshold voltage. The trigger signal disables
driver at time t2. The trigger drops down to LOW level in
time t3, but IC waits for complete minimum off−time.
Minimum off−time execution is blocked until CS pin
voltage goes above VTH_CS_RESET threshold. Next cycle
starts in time t6. The TRIG/DIS is low so driver is enabled
and ready to be turned on when VDS falls below VTH_CS_ON
threshold voltage thus the driver is turned on at time t6. The
trigger signal rises up to HIGH level at time t7, consequently
DRV turns−off and IC waits for high CS voltage to start
minimum off−time execution.
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VDS =V
CS
VTH _CS _ RESET
VTH _CS _ OFF
VTH _CS _ON
VTRIG/DIS
Min ON−time
VDRV
t
t0 t1 t2 t3 t4
Power
consumption
tDIS_TIM
Figure 58. Trigger Input Functionality Waveforms − Disable Mode Activation
In Figure 58 above, at t2 the CS pin rises to VTH_CS_OFF
and the SR MOSFET is turned−off. At t3 the TRIG/DIS
signal is held high for more than tDIS_TIM. NCP4305 enters
disable mode after tDIS_TIM. Driver output is disabled in
disable mode. The DRV stays low (disabled) during
transition to disable mode. Figure 59 shows disable mode
transition 2nd case − i.e. when trigger rising edge comes
during the trigger blank period. Figure 60 shows entering
into disable mode and back to normal sequences.
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VDS =V
CS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
Min ON−time
VDRV
t
t0 t1 t2 t3
Power
consumption
tDIS_TIM
tTRIGBLANK
Figure 59. Trigger Input Functionality Waveforms − Disable Mode Clock Initiation
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VDS =V
CS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
VDRV
t
t0 t1 t2 t3
Power
consumption
tDIS_TIM
tDIS_REC
t4
Disable mode
Min OFFtime
Figure 60. Trigger Input Functionality Waveforms − Disable and Normal Modes
Figures 61 and 62 shows exit from disable mode in detail.
NCP4305 requires up to tDIS_REC to recover all internal
circuitry to normal operation mode when recovering from
disable mode. The driver is then enabled after complete
tMIN_TOFF period when CS(VDS) voltage is over
VTH_CS_RESET threshold. Driver turns−on in the next cycle
on CS (VDS) falling edge signal only (t5 − Figure 61). The
DRV stays low during recovery time period. Trigger input
has to be low at least for tDIS_END time to end disable mode
and start with recovery. Trigger can go back high after
tDIS_END without recovery interruption.
/_ IIIIII W» H
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VDS =V
CS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
VDRV
t
t0 t1 t2 t3
Power
consumption
tDIS_REC
t4
Disable mode
Min OFFtime
Normal mode
Waits for
complete
tMIN_TOFF
t5 t8t6 t7
Figure 61. Trigger Input Functionality Waveforms − Exit from Disable Mode before the Falling Edge of the CS
Signal
—— —I—+————— ———I———+——+——— ——— www.0nsemi.com
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VDS =V
CS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
VDRV
t
t0 t1 t2 t3
Power
consumption
tDIS_END
t4
Min OFF−time
Normal mode
Waits for
complete
tMIN_TOFF
t5
Figure 62. Trigger Input Functionality Waveforms
time
Recovery
Disable
mode
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Figure 63. Trigger Input Functionality Waveforms
VDS
= VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
VDRV
t
t0 t1 t2 t3
Power
consumption
t4
Disable
Min OFF−time
Recovery Normal mode
Waits for
complete
tMIN_TOFF
t5
tDIS_REC
tMIN_TOFF
t6
mode
Figure 63 shows detail IC behavior after disable mode is
ended. The trigger pin voltage goes low at t1 and after
tDIS_REC IC leaves disable mode (t2). VDS voltage goes high
again at time t3 and this event starts minimum off−time timer
execution. Next VDS falling edge below VTH_CS_ON level
activates driver.
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VDS =V
CS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
VDRV
t
t0 t1 t2 t3
Power
consumption
t4
Min OFF−time
Recovery Normal mode
Waits for
complete
tMIN_TOFF
t5
tDIS_REC
tMIN_TOFF
t6 t7 t8
Figure 64. Trigger Input Functionality Waveforms
Disable
mode
Different situation of leaving from disable mode is shown
at Figure 64. Minimum off−time execution starts at time t2,
but before time elapses VDS voltage falls to negative
voltage. This interrupts minimum off−time execution and
the IC waits to another time when VDS voltage is positive
and then is again started the minimum off−time timer. The
IC returns into normal mode after whole minimum off−time
elapses.
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Figure 65. NCP4305 Operation after Start−Up Event
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VCCON
Min OFF− time
VDRV
VCC
Min ON−time
tMIN_TOFF tMIN_TOFF
tMIN_TON
Not complete
tMIN_TOFF
>IC
is not activated Complete
tMIN_TOFF
activates IC
tMIN_TOFF is stopped
due to VDS drops
below VTH_CS_RESET
t1t2 t3t4 t5t6t7 t8 t9 t10 t11t12
t13
t14
t15
Start−up event waveforms are shown at Figure 65. A
start−up event is very similar to an exit from disable mode
event. The IC waits for a complete minimum off−time event
(CS pin voltage is higher than VTH_CS_RESET) until drive
pulses can continue. Figure 65 shows how the minimum
off−time timer is reset when CS voltage is oscillating
through VTH_CS_RESET level. The NCP4305 starts
operation at time t1 (time t1 can be seen as a wake−up event
from the disable mode through TRIG/DIS or LLD pin).
Internal logic waits for one complete minimum off−time
period to expire before the NCP4305 can activate the driver
after a start−up or wake−up event. The minimum off−time
timer starts to run at time t1, because VCS is higher than
VTH_CS_RESET. The timer is then reset, before its set
minimum off−time period expires, at time t2 thanks to CS
voltage lower than VTH_CS_RESET threshold. The
aforementioned reset situation can be seen again at time t3,
t4, t5 and t6. A complete minimum off−time period elapses
between times t7 and t8 allowing the IC to activate a driver
output after time t8.
The NCP4305 works very well in CCM application
without any triggering method, but using some may improve
overall operation. Typical application schematics of CCM
flyback converters using two different primary triggering
techniques can be seen in Figures 66 and 67. Both provided
methods reduce the commutation losses and the SR
MOSFET drain voltage spike, which results in improved
efficiency.
P. d - 7E L
NCP4305
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Figure 66. Primary Triggering in Deep CCM Application Using Auxiliary Winding − NCP4305A, B, C or D
+
+
+
Vbulk
FLYBACK
CONTROL
CIRCUITRY
+Vout
GND
OK1
R5
R6
R7
R9
R10
C2 C3
C4
C5
C6
C7
D3
D4
D5
TR1
TR2
M1
NCP4305
R11
D6
D1
R1
M2
R12
R13
C8
D7
R14
VCC
DRV
FB CS
The application shown in Figure 66 is simplest and the
most cost effective solution for primary SR triggering. This
method uses auxiliary winding made of triple insulated wire
placed close to the primary winding section. This auxiliary
winding provides information about primary turn−on event
to the SR controller before the secondary winding reverses.
This is possible thanks to the leakage between primary and
secondary windings that creates natural delay in energy
transfer. This technique provides approximately 0.5%
efficiency improvement when the application is operated in
deep CCM and a transformer that has a leakage of 1% of
primary inductance is used.
Figure 67. Primary Triggering in Deep CCM Application Using Trigger Transformer − NCP4305A, B, C or D
Application from Figure 67 uses an ultra−small trigger
transformer to transfer primary turn−on information directly
from the primary controller driver pin to the SR controller
trigger input. Because the trigger input is rising edge
sensitive, it is not necessary to transmit the entire primary
driver pulse to the secondary. The coupling capacitor C5 is
used to allow the trigger transformers core to reset and also
to prepare a needle pulse (a pulse with width shorter than
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NCP4305
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37
100 ns) to be transmitted to the NCP4305 trigger input. The
advantage of needle trigger pulse usage is that the required
volt−second product of the pulse transformer is very low and
that allows the designer to use very small and cheap
magnetic. The trigger transformer can even be prepared on
a small toroidal ferrite core with outer diameter of 4 mm and
four turns for primary and secondary windings to assure
Lprimary = Lsecondary > 10 mH. Proper safety insulation
between primary and secondary sides can be easily assured
by using triple insulated wire for one or, better, both
windings.
This primary triggering technique provides
approximately 0.5% efficiency improvement when the
application is operated in deep CCM and transformer with
leakage of 1% of primary inductance is used.
It is also possible to use capacitive coupling (use
additional capacitor with safety insulation) between the
primary and secondary to transmit the trigger signal. We do
not recommend this technique as the parasitic capacitive
currents between primary and secondary may affect the
trigger signal and thus overall system functionality.
Minimum tON and tOFF Adjustment
The NCP4305 offers an adjustable minimum on−time and
off−time blanking periods that ease the implementation of a
synchronous rectification system in any SMPS topology.
These timers avoid false triggering on the CS input after the
MOSFET is turned on or off.
The adjustment of minimum tON and tOFF periods are
done based on an internal timing capacitance and external
resistors connected to the GND pin − refer to Figure 68 for
a better understanding.
Figure 68. Internal Connection of the MIN_TON Generator (the MIN_TOFF Works in the Same Way)
Current through the MIN_TON adjust resistor can be
calculated as:
IR_MIN_TON +
Vref
RTon_min
(eq. 5)
If the internal current mirror creates the same current
through RMIN_TON as used the internal timing capacitor (Ct)
charging, then the minimum on−time duration can be
calculated using this equation.
t
MIN_TON +Ct
Vref
IR_MIN_TON +Ct
Vref
Vref
R
MIN_TON
+Ct@RMIN_TO
N
(eq. 6)
The internal capacitor size would be too large if
IR_MIN_TON was used. The internal current mirror uses a
proportional current, given by the internal current mirror
ratio. One can then calculate the MIN_TON and
MIN_TOFF blanking periods using below equations:
tMIN_TON +1.00 * 10−4 *R
MIN_TON [ms] (eq. 7)
tMIN_TOFF +1.00 * 10−4 *R
MIN_TOFF [ms] (eq. 8)
Note that the internal timing comparator delay affects the
accuracy of Equations 7 and 8 when MIN_TON/
MIN_TOFF times are selected near to their minimum
possible values. Please refer to Figures 69 and 70 for
measured minimum on and off time charts.
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NCP4305
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Figure 69. MIN_TON Adjust Characteristics
Figure 70. MIN_TOFF Adjust Characteristics
RMIN_TON (kW)
906050403020100
0
1
2
4
5
6
7
10
tMIN_TON (ms)
100
3
8070
RMIN_TOFF (kW)
906050403020100
0
1
2
4
5
6
7
10
tMIN_TOFF (ms)
100
3
8070
8
9
8
9
The absolute minimum tON duration is internally clamped
to 55 ns and minimum tOFF duration to 245 ns in order to
prevent any potential issues with the MIN_TON and/or
MIN_TOFF pins being shorted to GND.
The NCP4305 features dedicated anti−ringing protection
system that is implemented with a MIN_TOFF blank
generator. The minimum off−time one−shot generator is
restarted in the case when the CS pin voltage crosses
VTH_CS_RESET threshold and MIN_TOFF period is active.
The total off-time blanking period is prolonged due to the
ringing in the application (refer to Figure 45).
Some applications may require adaptive minimum on and
off time blanking periods. With NCP4305 it is possible to
modulate blanking periods by using an external NPN
transistor − refer to Figure 71. The modulation signal can be
derived based on the load current, feedback regulator
voltage or other application parameter.
Figure 71. Possible Connection for MIN_TON and MIN_TOFF Modulation
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Maximum tON adjustment
The NCP4305Q offers an adjustable maximum on−time
(like the min_tON and min_tOFF settings shown above) that
can be very useful for QR controllers at high loads. Under
high load conditions the QR controller can operate in CCM
thanks to this feature. The NCP4305Q version has the ability
to turn−off the DRV signal to the SR MOSFET before the
secondary side current reaches zero. The DRV signal from
the NCP4305Q can be fed to the primary side through a
pulse transformer (see Figure 4 for detail) to a transistor on
the primary side to emulate a ZCD event before an actual
ZCD event occurs. This feature helps to keep the minimum
switching frequency up so that there is better energy transfer
through the transformer (a smaller transformer core can be
used). Also another advantage is that the IC controls the SR
MOSFET and turns off from secondary side before the
primary side is turned on in CCM to ensure no cross
conduction. By controlling the SR MOSFET’s turn off
before the primary side turn off, producing a zero cross
conduction operation, this will improve efficiency.
The Internal connection of the MAX_TON feature is
shown in Figure 72. Figure 72 shows a method that allows
for a modification of the maximum on−time according to
output voltage. At a lower VOUT, caused by hard overload
or at startup, the maximum on−time should be longer than at
nominal voltage. Resistor RA can be used to modulate
maximum on−time according to VOUT or any other
parameter.
The operational waveforms at heavy load in QR type
SMPS are shown in Figure 73. After tMAX_TON time is
exceeded, the synchronous switch is turned off and the
secondary current is conducted by the diode. Information
about turned off SR MOSFET is transferred by the DRV pin
through a small pulse transformer to the primary side where
it acts on the ZCD detection circuit to allow the primary
switch to be turned on. Secondary side current disappears
before the primary switch is turned on without a possibility
of cross current condition.
Figure 72. Internal Connection of the MAX_TON Generator, NCP4305Q
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VDS =V
CS
VTH_CS _RESET –(R
SHIFT _CS*ICS)
VTH_CS_OFF –(R
SHIFT _CS*ICS)
VTH_CS_ON–(R
SHIFT _CS*ICS)
VDRV
Min ON−time
t
Min OFFtime
tMIN_TON
tMIN_TOFF
ISEC
The tMIN _TON and tMIN_TOFF are adjustable by RMIN_TON and RMIN_TOFF resistors, tMAX_TON is adjustable by RMAX_TON
Turnon delay Turn −off delay
Primary virtual ZCD
detection delay
Max ONtime tMAX _TON
Figure 73. Function of MAX_TON Generator in Heavy Load Condition
Adaptive Gate Driver Clamp and automatic Light Load
Turn−off
As synchronous rectification system significantly
improves efficiency in most of SMPS applications during
medium or full load conditions. However, as the load
reduces into light or no−load conditions the SR MOSFET
driving losses and SR controller consumption become more
critical. The NCP4305 offers two key features that help to
optimize application efficiency under light load and no load
conditions:
1stThe driver clamp voltage is modulated and follows
the output load condition. When the output load decreases
the driver clamp voltage decreases as well. Under heavy
load conditions the SR MOSFET’s gate needs to be driven
very hard to optimize the performance and reduce
conduction losses. During light load conditions it is not as
critical to drive the SR MOSFET’s channel into such a low
RDSON state. This adaptive gate clamp technique helps to
optimize efficiency during light load conditions especially
in LLC applications where the SR MOSFETs with high
input capacitance are used.
Driver voltage modulation improves the system behavior
when SR controller state is changed in and out of normal or
disable modes. Soft transient between drop at body diode
and drop at MOSFET’s RDS(on) only improves stability
during load transients.
2nd − In extremely low load conditions or no load
conditions the NCP4305 fully disables driver output and
reduces the internal power consumption when output load
drops below the level where skip−mode takes place.
Both features are controlled by voltage at LLD pin. The
LLD pin voltage characteristic is shown in Figure 74. Driver
voltage clamp is a linear function of the voltage difference
between the VCC and LLD pins from VLLD_REC point up to
VLLD_MAX. A disable mode is available, where the IC
current consumption is dramatically reduced, when the
difference of VCC − VLLD voltage drops below VLLD_DIS.
When the voltage difference between the VCC − VLLD pins
increase above VLLC_REC the disable mode ends and the IC
regains normal operation. It should be noted that there are
also some time delays to enter and exit from the disable
mode. Time waveforms are shown at Figure 75. There is a
time, tLLD_DISH, that the logic ignores changes from disable
mode to normal or reversely. There is also some time
tLLD_DIS_R that is needed after an exit from the disable mode
to assure proper internal block biasing before SR controller
starts work normally.
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VDRVCLAMP
VCC −VLLD
ICC
VDRVMAX
VLLD_MAX
VLLD_DIS VLLD_REC
Figure 74. LLD Voltage to Driver Clamp and Current Consumption Characteristic (DRV Unloaded)
Figure 75. LLD Pin Disable Behavior in Time Domain
ICC
VCC−VLLD
DISABLE MODE NORMAL
NORMAL
NORMAL
DISABLE MODE
tLLD_DISH tLLD_DISH tLLD_DISH tLLD_DISH
VLLD_DIS
VLLD_REC
t
tLLD_DIS_R tLLD_DIS_R
The two main SMPS applications that are using
synchronous rectification systems today are flyback and
LLC topologies. Different light load detection techniques
are used in NCP4305 controller to reflect differences in
operation of both mentioned applications.
Detail of the light load detection implementation
technique used in NCP4305 in flyback topologies is
displayed at Figure 76. Using a simple and cost effective
peak detector implemented with a diode D1, resistors R1
through R3 and capacitors C2 and C3, the load level can be
sensed. Output voltage of this detector on the LLD pin is
referenced to controller VCC with an internal differential
amplifier in NCP4305. The output of the differential
amplifier is then used in two places. First the output is used
in the driver block for gate drive clamp voltage adjustment.
Next, the output signal is evaluated by a no−load detection
comparator that activates IC disable mode in case the load
is disconnected from the application output.
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Figure 76. NCP4305 Light Load and No Load Detection Principle in Flyback Topologies
Operational waveforms related to the flyback LLD
circuitry are provided in Figure 77. The SR MOSFET drain
voltage drops to ~ 0 V when ISEC current is flowing. When
the SR MOSFET is conducting the capacitor C2 charges−up,
causing the difference between the LLD pin and VCC pin to
increase, and drop the LLD pin voltage. As the load
decreases the secondary side currents flows for a shorter a
shorter time. C2 has less time to accumulate charge and the
voltage on the C2 decreases, because it is discharged by R2
and R3. This smaller voltage on C2 will cause the LLD pin
voltage to increase towards VCC and the difference between
LLD and VCC will go to zero. The output voltage then
directly reduces DRV clamp voltage down from its
maximum level. The DRV is then fully disabled when IC
enters disable mode. The IC exits from disable mode when
difference between LLD voltage and VCC increases over
VLLD_REC. Resistors R2 and R3 are also used for voltage
level adjustment and with capacitor C3 form low pass filter
that filters relatively high speed ripple at C2. This low pass
filter also reduces speed of state change of the SR controller
from normal to disable mode or reversely. Time constant
should be higher than feedback loop time constant to keep
whole system stable.
Figure 77. NCP4305 Driver Clamp Modulation Waveforms in Flyback Application Entering into Light/No Load
Condition
ISEC
VC2
VDRV
VC3
VLLDMAX
VLLD_REC
VLLD_DIS VDRVMAX
t
IC enters
disable mode
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Figure 78. NCP4305 Driver Clamp Modulation Circuitry Transfer Characteristic in Flyback Application
IOUT
VCC−VLLD
VDRV
IC enters
disable mode
VLLDMAX
VLLD_REC VLLD_DIS
VDRVMAX
t
The technique used for LLD detection in LLC is similar
to the LLD detection method used in a flyback with the exception the D1 and D2 OR−ing diodes are used to measure
the total duty cycle to see if it is operating in skip mode.
Figure 79. NCP4305 Light Load Detection in LLC Topology
The driver clamp modulation waveforms of NCP4305 in
LLC are provided in Figure 80. The driver clamp voltage
clips to its maximum level when LLC operates in normal
mode. When the LLC starts to operate in skip mode the
driver clamp voltage begins to decrease. The specific output
current level is determined by skip duty cycle and detection
circuit consists of R1, R2, R3, C2, C3 and diodes D1, D2.
The NCP4305 enters disable mode in low load condition,
when VCC−VLLD drops below VLLD_DIS (0.9 V). Disable
mode ends when this voltage increase above VLLD_REC
(1.0 V) Figure 81 shows how LLD voltage modulates the
driver output voltage clamp.
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VCS1
VCS2
VC2
VCCVLLD
DRV clamp
Skip operationNormal operation
(VC3)
IC enters
disable mode
VDRVMAX
VLLDMAX
VLLD_REC VLLD_DIS
t
Figure 80. NCP4305 Driver Clamp Modulation Waveforms in LLC Application
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VCC−V LLD
IOUT
DRV clamp
IC enters
disable mode
VLLDMAX
VLLD_REC VLLD_DIS
VDRVMAX
t
Figure 81. NCP4305 Driver Clamp Modulation Circuitry Characteristic in LLC Application
There exist some LLC applications where behavior
described above is not the best choice. These applications
transfer significant portion of energy in a few first pulses in
skip burst. It is good to keep SR fully working during skip
mode to improve efficiency. There can be still saved some
energy using LLD function by activation disable mode
between skip bursts. Simplified schematic for this LLD
behavior is shown in Figure 46. Operation waveforms for
this option are provided in Figure 83. Capacitor C2 is
charged to maximum voltage when LLC is switching. When
there is no switching in skip, capacitor C2 is discharged by
R2 and when LLD voltage referenced to VCC falls below
VLLD_DIS IC enters disable mode. Disable mode is ended
when LLC starts switching.
Figure 82. NCP4305 Light Load Detection in LLC Application − Other Option
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VCS1
VCS2
VC2
VCC−VLLD
DRV clamp
Skip operation
Normal operation
IC enters
disable mode
VDRVMAX
VLLDMAX
VLLD_REC
VLLD_DIS
t
Figure 83. NCP4305 Light Load Detection Behavior in LLC Application – Other Option
Power Dissipation Calculation
It is important to consider the power dissipation in the
MOSFET driver of a SR system. If no external gate resistor
is used and the internal gate resistance of the MOSFET is
very low, nearly all energy losses related to gate charge are
dissipated in the driver. Thus it is necessary to check the SR
driver power losses in the target application to avoid over
temperature and to optimize efficiency.
In SR systems the body diode of the SR MOSFET starts
conducting before SR MOSFET is turned−on, because there
is some delay from VTH_CS_ON detect to turn−on the driver.
On the other hand, the SR MOSFET turn off process always
starts before the drain to source voltage rises up
significantly. Therefore, the MOSFET switch always
operates under Zero Voltage Switching (ZVS) conditions
when in a synchronous rectification system.
The following steps show how to approximately calculate
the power dissipation and DIE temperature of the NCP4305
controller. Note that real results can vary due to the effects
of the PCB layout on the thermal resistance.
Step 1 − MOSFET Gate−to Source Capacitance:
During ZVS operation the gate to drain capacitance does
not have a Miller effect like in hard switching systems
because the drain to source voltage does not change (or its
change is negligible).
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Figure 84. Typical MOSFET Capacitances
Dependency on VDS and VGS Voltages
Ciss +Cgs )Cgd
Crss +Cgd
Coss +Cds )Cgd
Therefore, the input capacitance of a MOSFET operating
in ZVS mode is given by the parallel combination of the gate
to source and gate to drain capacitances (i.e. Ciss capacitance
for given gate to source voltage). The total gate charge,
Qg_total, of most MOSFETs on the market is defined for hard
switching conditions. In order to accurately calculate the
driving losses in a SR system, it is necessary to determine the
gate charge of the MOSFET for operation specifically in a
ZVS system. Some manufacturers define this parameter as
Qg_ZVS. Unfortunately, most datasheets do not provide this
data. If the Ciss (or Qg_ZVS) parameter is not available then
it will need to be measured. Please note that the input
capacitance is not linear (as shown Figure 84) and it needs
to be characterized for a given gate voltage clamp level.
Step 2 − Gate Drive Losses Calculation:
Gate drive losses are affected by the gate driver clamp
voltage. Gate driver clamp voltage selection depends on the
type of MOSFET used (threshold voltage versus channel
resistance). The total power losses (driving loses and
conduction losses) should be considered when selecting the
gate driver clamp voltage. Most of today’s MOSFETs for SR
systems feature low RDS(on) for 5 V VGS voltage. The
NCP4305 offers both a 5 V gate clamp and a 10 V gate
clamp for those MOSFET that require higher gate to source
voltage.
The total driving loss can be calculated using the selected
gate driver clamp voltage and the input capacitance of the
MOSFET:
PDRV_total +VCC @VCLAMP @Cg_ZVS @fSW (eq. 9)
Where:
VCC is the NCP4305 supply voltage
VCLAMP is the driver clamp voltage
Cg_ZVS is the gate to source capacitance of the
MOSFET in ZVS mode
fsw is the switching frequency of the target
application
The total driving power loss won’t only be dissipated in
the IC, but also in external resistances like the external gate
resistor (if used) and the MOSFET internal gate resistance
(Figure 50). Because NCP4305 features a clamped driver,
it’s high side portion can be modeled as a regular driver
switch with equivalent resistance and a series voltage
source. The low side driver switch resistance does not drop
immediately at turn−off, thus it is necessary to use an
equivalent value (RDRV_SIN_EQ) for calculations. This
method simplifies power losses calculations and still
provides acceptable accuracy. Internal driver power
dissipation can then be calculated using Equation 10:
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Figure 85. Equivalent Schematic of Gate Drive Circuitry
PDRV_IC +1
2@Cg_ZVS @VCLAMP 2@fSW @ǒRDRV_SINK_EQ
RDRV_SINK_EQ )RG_EXT )Rg_intǓ)Cg_ZVS @VCLAMP @fSW @ǒVCC *VCLAMPǓ
)1
2@Cg_ZVS @VCLAMP 2@fSW @ǒRDRV_SOURCE_EQ
RDRV_SOURCE_EQ )RG_EXT )Rg_intǓ
(eq. 10)
Where:
RDRV_SINK_EQ is the NCP4305x driver low side switch
equivalent resistance (0.5 W)
RDRV_SOURCE_EQ is the NCP4305x driver high side switch
equivalent resistance (1.2 W)
RG_EXT is the external gate resistor (if used)
Rg_int is the internal gate resistance of the
MOSFET
Step 3 − IC Consumption Calculation:
In this step, power dissipation related to the internal IC
consumption is calculated. This power loss is given by the
ICC current and the IC supply voltage. The ICC current
depends on switching frequency and also on the selected min
tON and tOFF periods because there is current flowing out
from the min tON and tOFF pins. The most accurate method
for calculating these losses is to measure the ICC current
when CDRV = 0 nF and the IC is switching at the target
frequency with given MIN_TON and MIN_TOFF adjust
resistors. IC consumption losses can be calculated as:
PCC +VCC @ICC (eq. 11)
Step 4 − IC Die Temperature Arise Calculation:
The die temperature can be calculated now that the total
internal power losses have been determined (driver losses
plus internal IC consumption losses). The package thermal
resistance is specified in the maximum ratings table for a
35 mm thin copper layer with no extra copper plates on any
pin (i.e. just 0.5 mm trace to each pin with standard soldering
points are used).
The DIE temperature is calculated as:
TDIE +ǒPDRV_IC )PCCǓ@RqJ−A )TA(eq. 12)
Where:
PDRV_IC is the IC driver internal power dissipation
PCC is the IC control internal power
dissipation
RqJA is the thermal resistance from junction to
ambient
TAis the ambient temperature
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PRODUCT OPTIONS
OPN Package UVLO [V] DRV clamp [V] Pin 5 function Usage
NCP4305ADR2G SOIC8 4.5 4.7 TRIG
LLC, CCM flyback, DCM flyback, forward,
QR, QR with primary side CCM control
NCP4305AMTTWG WDFN8 4.5 4.7 TRIG
NCP4305DDR2G SOIC8 4.5 9.5 TRIG
NCP4305DMNTWG DFN8 4.5 9.5 TRIG
NCP4305DMTTWG WDFN8 4.5 9.5 TRIG
NCP4305QDR2G SOIC8 4.5 9.5 MAX_TON QR with forced CCM from secondary side
ORDERING INFORMATION
Device Package Package marking Packing Shipping
NCP4305ADR2G SOIC8 NCP4305A SOIC−8
(Pb−Free) 2500 /Tape & Reel
NCP4305DDR2G NCP4305D
NCP4305QDR2G NCP4305Q
NCP4305AMTTWG WDFN8 5A WDFN−8
(Pb−Free) 3000 /Tape & Reel
NCP4305DMTTWG 5D
NCP4305DMNTWG DFN8 4305D DFN−8
(Pb−Free) 4000 /Tape & Reel
For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging
Specifications Brochure, BRD8011/D.
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PACKAGE DIMENSIONS
SOIC−8 NB
CASE 751−07
ISSUE AK
SEATING
PLANE
1
4
58
N
J
X 45_
K
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSION A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
6. 751−01 THRU 751−06 ARE OBSOLETE. NEW
STANDARD IS 751−07.
A
BS
D
H
C
0.10 (0.004)
DIM
AMIN MAX MIN MAX
INCHES
4.80 5.00 0.189 0.197
MILLIMETERS
B3.80 4.00 0.150 0.157
C1.35 1.75 0.053 0.069
D0.33 0.51 0.013 0.020
G1.27 BSC 0.050 BSC
H0.10 0.25 0.004 0.010
J0.19 0.25 0.007 0.010
K0.40 1.27 0.016 0.050
M0 8 0 8
N0.25 0.50 0.010 0.020
S5.80 6.20 0.228 0.244
−X−
−Y−
G
M
Y
M
0.25 (0.010)
−Z−
Y
M
0.25 (0.010) ZSXS
M
____
1.52
0.060
7.0
0.275
0.6
0.024
1.270
0.050
4.0
0.155
ǒmm
inchesǓ
SCALE 6:1
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
SOLDERING FOOTPRINT*
7mg szn Cu t DW‘ENS‘ONS AND TOLERANCING PER AsME VMSM. mm coNTRomNG DW‘ENS‘ON M‘LLIMETERS DW‘ENS‘ON b APPUES TO PLAIED TEPMWALAND \s MEASURED BEIWEEN a ‘5 AND a 3mm EPoM TEPMWAL UP a coPLANAPw APPUES TO THE EXPOSED PAD As WELL AS THE TERMWALS 5 DETNLS A AND 3 SHOW OPTIONAL coNsTRucnoNs FOR TEPMWAL: MILLIMETERS Dm MIN MAX A a an ‘ an A1 a an a as A3 a 2n REF a 25 \ a 35 SIDE VIEW mass m x 22‘ mass 209 \ 239 usnasc Dan E, an MD r-xnflmgan- 41$ BOTTOM VIEW www onsem com 51
NCP4305
www.onsemi.com
51
PACKAGE DIMENSIONS
DFN8 4x4
CASE 488AF
ISSUE C
NOTES:
1. DIMENSIONS AND TOLERANCING PER
ASME Y14.5M, 1994.
2. CONTROLLING DIMENSION: MILLIMETERS.
3. DIMENSION b APPLIES TO PLATED
TERMINAL AND IS MEASURED BETWEEN
0.15 AND 0.30MM FROM TERMINAL TIP.
4. COPLANARITY APPLIES TO THE EXPOSED
PAD AS WELL AS THE TERMINALS.
5. DETAILS A AND B SHOW OPTIONAL
CONSTRUCTIONS FOR TERMINALS.
DIM MIN MAX
MILLIMETERS
A0.80 1.00
A1 0.00 0.05
A3 0.20 REF
b0.25 0.35
D4.00 BSC
D2 1.91 2.21
E4.00 BSC
E2 2.09 2.39
e0.80 BSC
K0.20
L0.30 0.50
DB
E
C0.15
A
C0.15
2X
2X TOP VIEW
SIDE VIEW
BOTTOM VIEW
C
A
(A3) A1
8X
SEATING
PLANE
C0.08
C0.10
e
8X L
K
E2
D2
b
NOTE 3
14
58 8X
0.10 C
0.05 C
AB
PIN ONE
REFERENCE
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
SOLDERING FOOTPRINT*
8X
0.63
2.21
2.39
8X
0.80
PITCH
4.30
0.35
L1
DETAIL A
L
OPTIONAL
CONSTRUCTIONS
A1
A3
L
DETAIL B
MOLD CMPDEXPOSED Cu
ALTERNATE
CONSTRUCTIONS
L1 −−− 0.15
DETAIL B
NOTE 4
DETAIL A
DIMENSIONS: MILLIMETERS
PACKAGE
OUTLINE
fiUi E l M SIDE VIE rm on: @3444 A1 A3 1+ pm ‘ [ti/i 7x WU j 7 7 + H H “ 45L; ax b 69 BOWOM VIEW C Q [LID C am WI 3-. i nzrznzuczx
NCP4305
www.onsemi.com
52
PACKAGE DIMENSIONS
C
A
SEATING
PLANE
D
E
0.10 C
A3
A
A1
0.10 C
WDFN8 2x2, 0.5P
CASE 511AT
ISSUE O
DIM
A
MIN MAX
MILLIMETERS
0.70 0.80
A1 0.00 0.05
A3 0.20 REF
b0.20 0.30
D
E
e
L
PIN ONE
REFERENCE
0.05 C
0.05 C
A0.10 C
NOTE 3
L2
e
b
B
4
88X
1
5
0.05 C
L1
2.00 BSC
2.00 BSC
0.50 BSC
0.40 0.60
--- 0.15
BOTTOM VIEW
L
7X
L1
DETAIL A
L
ALTERNATE TERMINAL
CONSTRUCTIONS
L
DETAIL B
MOLD CMPDEXPOSED Cu
ALTERNATE
CONSTRUCTIONS
DETAIL B
DETAIL A
L2 0.50 0.70
B
TOP VIEW
SIDE VIEW
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ASME Y14.5M, 1994.
2. CONTROLLING DIMENSION: MILLIMETERS.
3. DIMENSION b APPLIES TO PLATED
TERMINAL AND IS MEASURED BETWEEN
0.15 AND 0.30 MM FROM TERMINAL TIP.
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
SOLDERING FOOTPRINT*
2.30
0.50
0.78
7X
DIMENSIONS: MILLIMETERS
0.30 PITCH
8X
1
PACKAGE
OUTLINE
RECOMMENDED
0.88
2X
2X
8X
e/2
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